Level-shifter, semiconductor integrated circuit, and control methods thereof

ABSTRACT

During a period corresponding to the former half of one cycle of a clock signal, a capacitor is charged to an intermediate potential between the respective precharged potentials of two level-shifters. Subsequently, during a period corresponding to the latter half of one cycle of the clock signal, the capacitor is connected to that one of the output nodes which shifts to a lower potential in the level-shifter on the upper stage, while a power source line is connected to the other output node which shifts to a higher potential. On the other hand, the capacitor is also connected to that one of the output nodes which shifts to the higher potential in the level-shifter on the lower stage, while the ground line is connected to the other output node which shifts to the lower potential. Consequently, there can be provided a semiconductor integrated circuit free from power dissipation that might have been caused by an internal power-source circuit. The semiconductor integrated circuit enables data transfer with a small amplitude and consumes an extremely small amount of current even when multi-bit data lines operate in parallel.

This is a continuation of application Ser. No. 08/382,530, filed Feb. 2,1995, now U.S. Pat. No. 5,581,506.

BACKGROUND OF THE INVENTION

The present invention relates to an improved level-shifter for use in amemory circuit in which a large number of data lines operate in parallelin synchronization with a clock signal on a semiconductor chip, such asan image memory, a synchronous dynamic random-access memory (SDRAM), ora static random-access memory (SRAM) or in a microprocessor forperforming parallel data processing, to a semiconductor integratedcircuit using the improved level-shifter, and to control methodsthereof.

A conventional level-shifter for shifting the amplitude level of aninput signal so as to produce an output signal of a different amplitudelevel is disclosed in Japanese Laid-Open Patent Publication No. 4-211515(by Nakagome et al. of Hitachi Ltd.). Below, a description will be givento the above prior art with reference to FIGS. 27 and 28.

FIG. 27(a) shows a clock-synchronization-type level-shifter and FIG.27(b) illustrates the operation thereof. In FIG. 27(a) are shown:synchronizing signals CLK(3) and XCLK(2) (clock signals); an inputsignal Vin(11) prior to level shifting; a High level power source VH(90)after level shifting; a Low level power source VL(91) after levelshifting; and a precharge power source VM(9) having an intermediatepower-source potential between the above High level potential VH and Lowlevel potential VL.

Next, the operation of the level-shifter will be described withreference to FIG. 27(b). When a PMOSFET(5) and an NMOSFET(6) are turnedON controlled by the clock signals CLK and XCLK, inverting operation isenabled so that the input Vin(11) is inverted and fetched. For example,if the Vin (11) is on the level of the power-source potential Vcc, theoutput has a Low level value represented by VL(91). Subsequently, whenthe above clock signals CLK and XCLK are inverted, the above PMOSFET(5)and NMOSFET(6) are turned OFF, while a PMOSFET(12) and an NMOSFET(9) areturned ON, so that the output is connected to the above precharge powersource VM(9) and precharged to the intermediate potential. If theforegoing operation, forming one cycle, is repeatedly performedafterwards, the above input Vin(11) is inverted in every cycle and theoutput is also inverted. In the foregoing operation, variations inoutput level become (VH-VM) and (VM-VL), which are smaller than avariation in input level of (Vcc-Vss). Briefly, the amount of chargerequired for charging or discharging the parasitic capacitance CD(10) ofthe output node Vout is satisfactorily reduced. Since the amount ofcharge is determined by the product of a variation in the potential ofthe capacitance and the capacitance value, if the ratio of the amplitudelevel of an input to that of the output becomes 1/10, e.g., due to thelevel-shifter, the amount of charge required for charging anddischarging the above output node also becomes 1/10. Compared with thecase where level shifting is not performed, a reduction in powerconsumption by one order of magnitude can be achieved.

Next, a generator of the above power sources VH(90) and VL(91) whichhave been shifted in level is shown in FIG. 28. The generator isdisposed on the same chip as the PMOSFET(6) and the like are disposed.As shown in the drawing, the generator has typically adopted a systemfor controlling the output transistor of a power-source circuit by acurrent-mirror-type output in which the potential obtained throughresistance division by means of resistors R1, R2, and R3 is used as areference potential. The voltage levels of the power sources VH(90) andVL(91) can arbitrarily be determined by adjusting the ratio among theabove resistances R1, R2, and R3.

In the above internal power-source circuit shown in FIG. 28, however, itis necessary to supply a power-source current at low resistance to adriver for driving a line with a large capacitance. Accordingly, theabove current mirror and output transistor increase in size so that theresistors R1, R2, R3, and R4 for generating the reference potentialcannot be composed of resistors with high resistance values. As aresult, the total sum of through currents IDC1 and IDC2 is increased tothe order of several milliamperes, so that the disadvantage of uselesslyincreased power consumption is caused.

If the above conventional disadvantage is considered in terms of powerconsumption, the power consumption of the level-shifter in FIG. 27becomes Ptoal=P1+P2. Here, the P2 is determined by the product of thecurrent consumed in driving the wiring capacitance with an amplitudeafter level shifting and an amplitude voltage after level shifting. Onthe other hand, the above P1 represents the power consumed uselessly bythe internal power-source circuit when it reduces voltage in generatingthe internal voltages VH and VL. The power consumption P1 is determinedby the product of the current consumed in driving the wiring capacitancewith the amplitude after level shifting and the amount of voltage(Vcc-VH+VL) reduced by the above internal power-source circuit. If theamount of the reduced voltage is large, i.e., if the output amplitudevalue is to be set smaller, the useless power consumption P2 is furtherincreased.

Moreover, since 64-bit data lines, 128-bit data lines, or 256-bit datalines operate in parallel on the same chip in an image memory or thelike, power consumption amounts to the value obtained by multiplying theabove-mentioned power consumption by the number of bits, which isconsiderably large as a whole.

In the level-shifter shown in FIG. 27(a), the delay time between thechange in the input Vin(11) caused in synchronization with the clocksignal and the production of the output is represented by td1 shown inFIG. 27(b). The delay time td1 is primarily determined by the timerequired by the input to become lower and higher than the abovepower-source potentials VH(90) and VL(91) by the threshold voltages ofthe above PMOSFET(4) and NMOSFET(7), respectively. The power-sourcepotentials VH(90) and VL(91) are the source potentials of the abovePMOSFET(4) and NMOSFET(7) serving as the input gates. However, if thevalue of the power-source potential VH(90) is further reduced and thevalue of the power-source potential VL(91) is further increased in orderto further reduce the output amplitude, the above PMOSFET(4) andNMOSFET(7) are turned ON only when the input approaches the lowerpotential Vss or the higher potential Vcc furthermore. Consequently, thedelay time td1 is disadvantageously increased.

To prevent power dissipation due to the above internal power-sourcecircuit, Japanese Laid-Open Patent Publication No. 4-302463 (byTakashima et el. of Toshiba Corporation) discloses a technique asillustrated in FIG. 29(a). With the technique, circuits 500 each showingan identical power-source current varying characteristic with respect toelapsed time, i.e., circuits 500 each having the same resistance in theON state are connected in series between the power source Vcc and theground line Vss so that the terminal voltage placed effectively on eachof the circuits becomes half the power-source voltage Vcc. In otherwords, by connecting in series the circuits having equal power-sourcecurrents at each point of elapsed time as shown in FIGS. 29(a) and29(b), i.e., by connecting in series the circuits having equal internalresistances in the ON state, each of the circuits 500 causes a voltagedrop, while performing its intrinsic circuit operation, therebyperforming the same function as performed by the above firstconventional internal power-source circuit.

However, the above prior art of FIG. 29 is disadvantageous in that, ifthe power-source currents for the respective circuits are not equal ateach point of elapsed time, the voltage determined by resistive divisionvaries, so that the effective terminal voltages of the respectivecircuits vary. Moreover, since the power-source current for each of thecircuits should be supplied by the ground current of its one-stage uppercircuit, the current cannot be reused if the condition as shown in FIG.29(b) is not satisfied, i.e., if the power-source currents I1 and I2 andground currents I1X and I2X of the respective circuits 500 are not equalat each point of elapsed time. If a consideration is given to the senseamplifying operation of simultaneously restoring the potentials ofindividual bit lines which consume a large current in a DRAM, e.g.,transistors with different functions are generally varied in size and inwiring resistance so that a time difference is automatically producedbetween charging operation accompanied by the flow of the power-sourcecurrent and discharging operation accompanied by the flow of the groundcurrent, thus preventing through currents from the power sources of therespective circuits to the ground. Therefore, it is impossible toequalize the charging currents and discharging currents of therespective circuits at each point of elapsed time without increasing thethrough currents. To cause the upper-stage and lower-stage circuits 500to operate identically, it is necessary to supply a current from anadditional circuit to the lower-stage circuit 500. Hence, it can beconcluded that the prior art is still disadvantageous in that ituselessly consumes more current than is needed to drive the wiringcapacitance.

SUMMARY OF THE INVENTION

An object of the present invention is to implement the higher-speedoperation of level-shifters with lower power consumption, irrespectiveof the number of the level-shifters to be operated simultaneously. Thelevel-shifter consumes only the current required to drive the wiringcapacitance on the output side thereof at a high speed so that, unlikethe prior art, useless power consumption is eliminated.

To attain the above object, the present invention has adopted astructure in which a charge is accumulated by means of a capacitor orthe like so that the accumulated charge is reused for a potential changeat the output nodes of the level-shifter, thereby eliminating uselessconsumption of power as consumed conventionally.

In a specific structure, the level-shifter of the present inventionreceives an input signal switching between two different values, whileoperating in synchronization with a clock signal, and shifts theamplitude value of the above input signal to another amplitude value sothat the resulting signal is outputted from a single output node, theabove level-shifter comprising: a first charge supplying means to beprecharged to a first potential; a second charge supplying means to beprecharged to a second potential which is different from the above firstpotential; and a discharging means for selecting either one of the abovecharge supplying means in accordance with the above input signal andreleasing the charge accumulated in the selected charge supplying meansto the above output node.

In another specific structure, the level-shifter of the presentinvention receives an input signal switching between two differentvalues, while operating in synchronization with a clock signal, andshifts the amplitude value of the above input signal to anotheramplitude value so that the resulting signal is outputted from a pair ofoutput nodes, the above level-shifter comprising: a first chargesupplying means to be precharged to a first potential; a second chargesupplying means to be precharged to a second potential which isdifferent from the above first potential; and a discharging means forreleasing, in accordance with the above input signal, the chargeaccumulated in the above first charge supplying means to either one ofthe above pair of output nodes, while releasing the charge accumulatedin the above second charge supplying means to the other output node.

Furthermore, the present invention is constituted so that the abovecharge supplying means is specified, the charge supplying means iscomposed of a capacitor, and the capacitance value of the abovecapacitor is determined so that the ratio of the capacitance value tothe parasitic capacitance value of the output node becomes equal to theratio of the difference between a desired potential at the output nodeand the second potential, which is the precharged potential at the aboveoutput node, to the difference between the first potential, which is theprecharged potential in the above capacitor, and the above potential atthe output node.

Additionally, in a specific structure, a semiconductor integratedcircuit of the present invention has a plurality of level-shifters eachof which receives an input signal switching between two differentvalues, while operating in synchronization with a clock signal, andshifts the amplitude value of the above input signal to anotheramplitude value so that the resulting signal is outputted from an outputnode, the above semiconductor integrated circuit comprising: a chargevariation equalizing means for causing any two of the abovelevel-shifters to present charge variations in which charges move in theopposite directions at the output nodes and the charges have equalabsolute values with respect to elapsed time; and a chargeredistributing means for causing the charges to move between the abovetwo level-shifters which present equal charge variations.

The present invention is also constituted so that the charge variationequalizing means consists of a precharging means for precharging twooutput nodes having substantially equal parasitic capacitance values toequal potentials, a charge accumulating means for accumulating a charge,and a charging means for charging the above charge accumulating means toa potential different from the above specified potential.

Furthermore, a method of controlling a semiconductor integrated circuitof the present invention comprises the steps of: during a first periodof one cycle of the clock signal, precharging the two output nodes ineach of the level-shifters to equal potentials; during a second periodof one cycle of the above clock signal, charging each of the respectivecharge accumulating means for the two level-shifters between which thecharge is redistributed to a specified potential between the twoprecharged potentials of the above two level-shifters between which thecharge is redistributed and during a third period after the above firstand second periods of one cycle of the above clock signal, subsequentlyconnecting one of the respective charge accumulating means of the abovetwo level-shifters between which the charge is redistributed to that oneof the output nodes which shifts to the higher potential in one of theabove two level-shifters between which the charge is redistributed,while connecting the other charge accumulating means to that one of theoutput nodes which shifts to the lower potential in the otherlevel-shifter.

With the above structures, in the level-shifter of the presentinvention, a charge is accumulated in the charge supplying means so thatthe accumulated charge is released to the output node, thereby varyingthe potential at the output node.

In this case, if the capacitance value of the charge supplying means(capacitor) equals to 1/10 of the parasitic capacitance value of theoutput node in the level-shifter, the output amplitude value at theoutput node is shifted in level to 1/10 of the input amplitude value.

Since the operation of the level-shifter is performed by discharging thecharge supplying means, it is no more necessary to provide an internalpower-source circuit as in a conventional embodiment, so that thecurrent consumed uselessly by a through current is not increased.

Moreover, in the two level-shifters connected in series of the presentinvention, the charge released by the output node which shifts to thelower potential in one level-shifter is reused to raise the potential atthe output node which shifts to the higher potential in the otherlevel-shifter, so that a further reduction in power consumption can beachieved accordingly.

Furthermore, according to the present invention, the charge released bythat one of the output nodes which shifts to the lower potential isaccumulated in the charge accumulating means, so that the accumulatedcharge is reused to raise the potential of that one of the output nodeswhich shifts to the higher potential.

The above objects and novel features of the present invention will bemore apparent from the reading of the following description inconjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings illustrate the preferred embodiments of thepresent invention.

In the drawings:

FIG. 1 is a view showing a level-shifter of a first embodiment;

FIG. 2 is a view illustrating the operation of the level-shifter of thefirst embodiment;

FIG. 3 is a view showing a level-shifter of a second embodiment;

FIG. 4 is a view showing a variation of the level-shifter of the secondembodiment;

FIG. 5 is a view illustrating the operation of the level-shifter of thesecond embodiment;

FIG. 6(a) shows another variation of the level-shifter of the secondembodiment, in which another charge supplying means is composed of apower-source line of a higher potential;

FIG. 6(b) shows still another variation of the level-shifter of thesecond embodiment, in which another charge supplying means is composedof a power-source line of a lower potential;

FIG. 7 is a view showing a semiconductor integrated circuit of a thirdembodiment;

FIG. 8 is a view illustrating the operation of the semiconductorintegrated circuit of the third embodiment;

FIG. 9 is a view showing a variation of the semiconductor integratedcircuit of the third embodiment, in which a capacitor is used in common;

FIG. 10 is a view showing a semiconductor integrated circuit of a fourthembodiment;

FIG. 11 is a view showing the controlling of switches in thesemiconductor integrated circuit of the fourth embodiment;

FIG. 12 is a view illustrating the operation of the semiconductorintegrated circuit of the fourth embodiment;

FIG. 13 is a view showing a variation of the semiconductor integratedcircuit of the fourth embodiment;

FIG. 14 is a view showing the controlling of switches in the variationof the semiconductor integrated circuit of the fourth embodiment;

FIG. 15 is a view illustrating the operation of the variation of thesemiconductor integrated circuit of the fourth embodiment;

FIG. 16 is a view showing a semiconductor integrated circuit of a fifthembodiment;

FIG. 17 is a view showing a circuit to be added to the semiconductorintegrated circuit of the fifth embodiment;

FIG. 18 is a view illustrating the operation of the semiconductorintegrated circuit of the fifth embodiment;

FIG. 19 is a view showing an equivalent circuit to the semiconductorintegrated circuit of the fifth embodiment;

FIG. 20 is a view showing the effect of the fifth embodiment;

FIG. 21 is a view showing the structure of an internal power-sourcecircuit in the fifth embodiment;

FIG. 22 is a view showing a semiconductor integrated circuit of a sixthembodiment;

FIG. 23 is a view showing a circuit to be added to the semiconductorintegrated circuit of the sixth embodiment;

FIG. 24 is a view illustrating the operation of the semiconductorintegrated circuit of the sixth embodiment;

FIG. 25 is a view showing a semiconductor integrated circuit of aseventh embodiment;

FIG. 26 is a view illustrating the operation of the semiconductorintegrated circuit of the seventh embodiment;

FIG. 27(a) is a view showing the structure of a first conventionallevel-shifter;

FIG. 27(b) is a view illustrating the operation of the firstconventional level-shifter;

FIG. 28 is a view showing the structure of a conventional internalpower-source circuit;

FIG. 29(a) is a view showing the structure of a second conventionallevel-shifter;

FIG. 29(b) is a view illustrating the operation of the secondconventional level-shifter;

FIG. 30 is a view showing an equivalent circuit to the secondconventional level-shifter;

FIG. 31 is a circuit diagram showing an eighth embodiment;

FIG. 32 is a block diagram showing a principal portion of alevel-shifter;

FIG. 33 is a view illustrating the operation of the level-shifter ofFIG. 32;

FIG. 34 is a view illustrating the case wherein the level-shifter areconnected in series;

FIG. 35 is a view showing an equivalent circuit to the circuit of FIG.31;

FIG. 36(a) is a view showing a first operating waveform of the eighthembodiment and a variation thereof;

FIG. 36(b) is a view showing a second operating waveform of the eighthembodiment and a variation thereof;

FIG. 37 is a circuit diagram showing a variation of the eighthembodiment;

FIG. 38 is a circuit diagram of an equivalent circuit to the circuit ofthe variation of the eighth embodiment;

FIG. 39 is a circuit diagram showing a ninth embodiment;

FIG. 40 is a view showing an operating waveform of the circuit of theninth embodiment;

FIG. 41 is a view illustrating a circuit of a tenth embodiment;

FIG. 42 is a view illustrating the operation of the circuit of the tenthembodiment;

FIG. 43(a) is a circuit diagram showing a charge redistributing meanscomposed of two MOSFETs;

FIG. 43(b) is a view illustrating the characteristic of the operationdelay time of the MOSFET;

FIG. 44 is a view showing a circuit of an eleventh embodiment;

FIG. 45 is a view showing the structure of a frequency divider;

FIG. 46 is a view showing operating frequencies of the individual groupsof the eleventh embodiment;

FIG. 47 is a view showing the effect of the eleventh embodiment;

FIG. 48 is a view showing a circuit of a twelfth embodiment;

FIG. 49 is a view illustrating the operation of the twelfth embodiment;

FIG. 50 is a view showing the operation of the twelfth embodiment andthe operation of the seventh embodiment for comparison;

FIG. 51 is a view showing a principal portion of a circuit of athirteenth embodiment;

FIG. 52 is a view showing the overall structure of the circuit of thethirteenth embodiment;

FIG. 53 is a view showing the internal structure of a potentialdetecting circuit composed of an N-type MOSFET;

FIG. 54 is a view showing the internal structure of potential detectingcircuit composed of a P-type MOSFET;

FIG. 55 is a view showing the operation delay characteristics of theN-type MOSFET and P-type MOSFET;

FIG. 56 is a view showing the overall structure of circuit of a firstvariation of the ninth embodiment;

FIG. 57 is a view illustrating the operation of the first variation ofthe ninth embodiment;

FIG. 58 is a view showing the overall structure of a circuit of a secondvariation of the ninth embodiment; and

FIG. 59 is a view illustrating the operation of the second variation ofthe ninth embodiment.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Below, the preferred embodiments of the present invention will bedescribed with reference to the accompanying drawings.

(First Embodiment)

FIG. 1 is a schematic view of a circuit according to a first embodimentof the present embodiment. FIG. 1 shows an inverter-type level-shifterwhich operates in synchronization with a clock signal. In the drawing,Vin represents an input, Vout represents an output node, referencenumerals 2 and 3 designate a complementary pair of clock signals CLK andKCLK, respectively, Vcc represents an external power-source line on thehigher potential side, Vss represents a power-source line on the lowerpotential side composed of an earth line, VM represents a power-sourceline of an intermediate potential ((Vcc-Vss)/2) between the above higherpotential Vcc and lower potential Vss.

A reference numeral 1a designates a first capacitor which functions as afirst charge supplying means for releasing the accumulated charge and areference numeral 1b designates a second capacitor which functions as asecond charge supplying means for releasing the accumulated charge,similarly to the first capacitor 1a.

A reference numeral 9 designates a discharging means consisting of twoPMOSFETs 4 and 5 and two NMOSFETs 6 and 7. A reference numeral 11designates a precharge circuit composed of one NMOSFET 8 and one PMOSFET12. Reference numerals 70 and 71 designate a PMOSFET and an NMOSFET,respectively, which connect the power-source lines Vcc and Vss to thecorresponding capacitors 1a and 1b in the ON state so that they arecharged.

The capacitance values CV of the above two capacitors 1a and 1b areequal to each other. The ratio of the capacitance value CV to theparasitic capacitance value of the output node Vout is set equal to theratio of the difference between a desired potential at the output nodeVout and the precharged potential (Vcc/2) at the above output node Voutto the difference between the first potentials (Vcc (e.g., 3.0 V) andVss (0 V)), which are the precharged potentials in the above capacitors1a and 1b, and the above desired potential at the output node Vout. Forexample, if the parasitic capacitance of the output is 3 pF, thecapacitance value CV of the capacitor is 0.3 pF.

Next, a description will be given to a control method of the abovelevel-shifter of FIG. 1 with reference to FIG. 2.

The input Vin is a signal switched between different two values as shownin FIG. 2. The PMOSFET 4 and NMOSFET 7 are controlled by the input Vin.The other PMOSFETs 5, 12, and 70 are controlled by the clock signalCLK(3) shown in FIG. 2, while the other NMOSFETs 6, 8, and 71 arecontrolled by the inverted clock signal XCLK(2) shown in FIG. 2.

That is, as shown in FIG. 2, during a period α (first period)corresponding to a half of one cycle of the clock signal CLK(3), thepower-source line VM of the intermediate potential is connected by thetwo MOSFETs 8 and 12 of the precharge circuit 11 to the output node Voutso that the output node Vout is precharged to the intermediatepotential. During the period 5, furthermore, the PMOSFET 5 and NMOSFET 6of the discharging means 9 are turned OFF, while the PMOSFET 70 andNMOSFET 71 are turned ON, thereby connecting the external power-sourceline Vcc on the higher potential side to the first capacitor 1a so thatit is charged to the higher potential and connecting the externalpower-source line Vss on the lower potential side to the secondcapacitor 1b so that it is charged to the lower potential. The chargingof the capacitors 1a and 1b is conducted during the above period α(coincident with the first period), but it may also be conducted duringa period (second period) different from the period during which theabove output node is precharged, provided that it is conducted at astage before the charges accumulated therein are outputted to the outputnodes Vout and XVout.

Subsequently, during a period β (third period) corresponding to theother half of one cycle of the above clock signal CLK(3) after theperiod a, the precharging operation by the precharge circuit 11 ishalted, while the above PMOSFET 70 and NMOSFET 71 are turned OFF,thereby halting the charging of the capacitors 1a and 1b. At the sametime, the PMOSFET 5 and NMOSFET 6 of the above discharge circuit 9 areturned ON. In the discharge circuit 9, either of the PMOSFET 5 andNMOSFET 6 is turned ON in accordance with the input Vin, so that thecharge accumulated in either of the capacitors 1a and 1b is released tothe output node Vout via the discharge circuit 9.

Thus, in the present embodiment, the voltage level at the output nodeVout is reduced to have a small amplitude by redistributing the chargefrom the capacitors 1a and 1b with no power dissipation. Consequently,it is no more necessary to provide a low power-source voltage inside thechip, which has been necessary in a conventional embodiment, so that anincrease in consumed current due to a through current in the case wherean internal power-source circuit is added and consumption of extra powerdue to a voltage drop in the internal power-source circuit can beprevented.

Moreover, since the above capacitors 1a and 1b maintain the sourcevoltages of the input MOSFETs 4 and 7 at a high value in the vicinity ofthe power-source voltage Vcc on the High side and at a low value in thevicinity of the ground voltage Vss on the Low side in the early periodof operation, the above input MOSFETs 4 and 7 are effectively turned ONwith ease. Consequently, the delay time td1 in the case of obtaining theoutput of small amplitude is no more increased as in the conventionalembodiment, so that it can be concluded that the present embodiment hasthe effect of reducing the delay time td2.

(Second Embodiment)

Next, a second embodiment of the present invention will be describedwith reference to FIG. 3.

The present embodiment shows an application of the level-shifter whichreceives a complementary pair of inputs and produces a complementarypair of outputs. In FIG. 3, Vin and XVin represent the complementarypair of inputs and Vout and XVout represent the complementary pair ofoutput nodes. A reference numeral 9' designates a discharging meansconsisting of two NMOSFETs 13 disposed on the input side and twoPMOSFETs 14. A reference numeral 11' designates a precharging meanscomposed of a PMOSFET 17 disposed on the output side. A referencenumeral 11' designates a precharging means composed of a PMOSFET 17disposed on the output side. Reference numerals 1a and 1b designatecapacitors serving as charge supplying means, similarly to the abovefirst embodiment. Reference numerals 19 and 20 designate an NMOSFET anda PMOSFET, respectively, while Vc represents a power-source line of aspecified potential and Vs represents a power-source line of anotherspecified potential lower than that of the above power-source line Vc.The power-source line Vs is composed of an earth line. The abovecapacitor 1a is charged to the potential of the power-source line Vc.The capacitor 1b is charged to the potential of the power-source lineVs.

The two NMOSFETs 13 of the above discharging means 9' are capable ofreleasing the charge in the capacitor 1a to the complementary pair ofoutput nodes Vout and XVout. The above two PMOSFET 14 are capable ofreleasing the charge in the capacitor 1b to the complementary pair ofoutput nodes Vout and XVout. The PMOSFET 17 of the above prechargingmeans 11' precharges the complementary pair of output nodes Vout andXVout to the same potential by short-circuiting them.

The above capacitors 1a and 1b have the same capacitance value CV. Thecapacitance value CV is set by equalizing the parasitic capacitancevalues of the above respective output nodes Vout and XVout so that theratio of the capacitance value CV to the parasitic capacitance value ofeach of the above output nodes Vout and XVout becomes equal to the ratioof the desired output amplitude value of each of the output nodes Voutand XVout to the difference between each of the potentials for chargingthe above capacitors 1a and 1b and the precharged potential at theoutput nodes Vout and XVout.

Next, a description will be given to a control method of thelevel-shifter of FIG. 3. Either one of the two NMOSFETs 13 on the inputside of the discharge circuit 9' and either one of the two PMOSFETs 14are controlled by either one of the complementary pair of inputs Vin andXVin, while the other NMOSFET 13 and the other PMOSFET 14 are controlledby the other of the complementary pair of inputs Vin and XVin. ThePMOSFET 17 of the precharge circuit 11' and the PMOSFET 20 arecontrolled by the clock signal CLK, while the NMOSFET 19 is controlledby the inverted clock signal XCLK. The above complementary pair of clocksignals CLK and KCLK and the complementary pair of inputs Vin and XVinare shown in FIG. 5.

Subsequently, a description will be given to the operation of the abovelevel-shifter of FIG. 3 with reference to FIG. 5.

In FIG. 5, during the period α corresponding to the former half of onecycle of the clock signal, i.e., during the period during which thecomplementary pair of inputs Vin and XVin have equal values, the PMOSFET17 of the precharge circuit 11' is turned ON so that the complementarypair of output nodes Vout and XVout are short-circuited and prechargedto equal potentials. During the period α, furthermore, the four MOSFETs13 and 14 of the discharge circuit 9' are turned OFF, while the NMOSFET19 and PMOSFET 20 are turned ON, so that the capacitor 1a is charged tothe lower potential of the power-source line Vs on the lower potentialside and the capacitor 1b is charged to the higher potential of thepower-source line Vc on the higher potential side.

Next, during the period β corresponding to the latter half of one cycleof the clock signal, i.e., during the period during which a specifiedpotential difference appears between the complementary pair of inputsVin and XVin, the PMOSFET 17 of the precharge circuit 11', the NMOSFET19, and the PMOSFET 20 are turned OFF, while the upper NMOSFET 13 andlower PMOSFET 14 of the discharge circuit 9' are turned ON, so that thecharge in the capacitor 1a charged to the lower potential is released tothe output node Vout and the charge in the capacitor 1b charged to thehigher potential is released to the output node XVout. As a result, theoutputs with the following potential difference (amplitude value) ΔVtherebetween are observed at the complementary pair of output nodes Voutand XVout:

    ΔV=2×Vcc (1+CD/CV)

where CD represents the parasitic capacitance value of each of thecomplementary pair of output nodes Vout and XVout.

Thus, in the present embodiment also, the voltage level of thecomplementary pair of output nodes Vout and XVout is reduced to have asmall amplitude by redistributing the charge from the capacitors 1a and1b with no power dissipation, similarly to the above first embodiment.Consequently, an increase in consumed current due to a through currentin the conventional case where the low power-source voltage is providedinside the chip or consumption of extra power due to a voltage drop inthe internal power-source circuit can be prevented.

(First Variation of Second Embodiment)

FIG. 4 shows a level-shifter which is a variation of the level-shifterof FIG. 3. The level-shifter of FIG. 4 is different from thelevel-shifter of FIG. 3 in that a single capacitor 1 is provided,instead of the capacitor 1a charged to the lower potential and thecapacitor 1b charged to the higher potential.

In the capacitor 1, one of a pair of plate electrodes is charged to thehigher potential of the power-source line Vc, while the other plateelectrode is charged to the lower potential of the power-source line Vs,so that, during the period β during which the specified potentialdifference appears between the above complementary pair of inputs Vinand XVin, the pair of plate electrodes are connected to thecomplementary pair of output nodes Vout and XVout, respectively.

Compared with the above second embodiment, the number of capacitors usedin the present variation can be reduced from two to one, so that theeffect of halving the layout area occupied by the capacitor can beachieved.

(Second Variation of Second Embodiment)

FIGS. 6(a) and 6(b) show a second variation of the above level-shifterof FIG. 3. The level-shifter of FIGS. 6(a) and 6(b) is different fromthe above level-shifter of FIG. 3 in that a single capacitor 1 (chargesupplying means) is provided, instead of the two capacitors 1a and 1b ascharge supplying means. Another charge supplying means is composed ofthe power-source line Vcc of the higher potential in FIG. 6(a), thoughit is composed of the power-source line Vss of the lower potential inFIG. 6(b). The level-shifter in the present variation is constituted sothat, if either one of the two NMOSFETs 13 of the discharging means 9'is turned ON in response to the input Vin or XVin, the charge on theabove power-source line Vcc of the higher potential and the charge onthe above power-source line Vss of the lower potential are releasedindividually to the complementary pair of outputs Vout and XVout via theabove NMOSFET 13 that has been turned ON. As for the other structure, itis the same as that of the level-shifter of FIG. 3, so that thedescription thereof will be omitted.

(Third Embodiment)

FIG. 7 shows a semiconductor integrated circuit according to a thirdembodiment of the present invention, in which the abovecomplementary-type level-shifter of FIG. 6(a) is placed on the upperstage and the complementary-type level-shifter of FIG. 6(b) is placed onthe lower stage. In the above level-shifter on the upper stage, twoinput gate MOSFETs 16 consist of PMOSFETs. In the above level-shifter onthe lower stage, two MOSFETs 15 consist of NMOSFETs. The MOSFET 18constituting the precharge circuit is composed of an NMOSFET.

The capacitors 1a and 1b are connected to the power-source line VM ofthe intermediate potential via the NMOSFETs 20a and 20b, respectively.The intermediate potential on the power-source line VM is between thehigher potential on the power-source line Vcc and the lower potential onthe power-source line Vss (VM=(Vcc-Vss)/2). The above two NMOSFETs 20aand 20b are controlled by the inverted clock signal KCLK. In thedrawing, a reference numeral 21 designates an NMOSFET for equalizing thepotentials of the two capacitors 1a and 1b, which is also controlled bythe inverted clock signal XCLK.

As for the other structure, it is the same as that of the structuresshown in FIG. 6(a) and 6(b), so that the description thereof will beomitted by providing like reference numerals.

Next, a description will be given to a control method of the abovesemiconductor integrated circuit.

As shown in FIG. 8, a complementary pair of inputs Xin1 to thelevel-shifter on the upper stage have sufficient potentials to turn ONthe input gate PMOSFET 16 and turn OFF the input gate PMOSFET 14completely, while a complementary pair of inputs Vin2 and XVin2 to thelevel-shifter on the lower stage have sufficient potentials to turn ONthe input gate PMOSFET 13 and turn OFF the input gate PMOSFET 15completely.

As shown in FIG. 8, during the period α corresponding to the former halfof one cycle of the clock signal, i.e., during the period during whichthe complementary pairs of inputs have equal values, a complementarypair of output nodes Vout1 and XVout1 of the level-shifter on the upperstage and a complementary pair of output nodes Vout2 and XVout2 of thelevel-shifter on the lower stage are equalized by turning ON the PMOSFET17 and by turning ON of the PMOSFET 18, respectively. During the periodα, furthermore, the turning ON of the two NMOSFETs 20a and 20b causesthe capacitors 1a and 1b to be precharged to the intermediate potentialby means of the power-source line VM of the intermediate potential.

Subsequently, during the period β corresponding to the latter half ofone cycle of the clock signal, i.e., during the period during which apotential difference is produced between each complementary pair ofinputs so that the input XVin1, e.g., is reduced to the lower potentialVss and the input Vin2 is increased to the higher potential Vcc, thepower-source line Vcc of the higher potential in the level-shifter onthe upper stage is connected to the output node XVout1 via the PMOSFET16 controlled by the above input XVin1 so that the output node XVout isincreased to the higher potential, while the output node Vout1 isconnected to the capacitor 1a via the PMOSFET 14 controlled by the aboveinput XVin1 so that an current Iu is allowed to flow from the outputnode Vout1 to the capacitor 1a.

During the above period β, on the other hand, the power-source line Vssof the lower potential in the level-shifter on the lower stage isconnected to the output node Vout2 via the NMOSFET 13 controlled by theabove input Vin2 so that the output node Vout2 is reduced to the lowerpotential Vss. The output node XVout2 is connected to the capacitor 1bvia the NMOSFET 15 controlled by the above input Vin2 so that a currentIL is allowed to flow from the capacitor 1b to the output node XVout2.As a result, the potential of the output node XVut2 is raised to a valuehigher than the precharged potential during the period α.

The charge accumulated by the current Iu flowing from the output nodeVout1 of the level-shifter on the upper stage is resultingly accumulatedin the output node XVout2 by the current IL flowing from the capacitor1b of the level-shifter on the lower stage, thereby reusing the charge.

Thus, in the present embodiment, the charge can be reused by the twolevel-shifters so that a further reduction in power consumption can beachieved.

The capacitance value CV of each of the capacitors 1a and 1b is set sothat the ratio of the capacitance value CV to the parasitic capacitancevalue CD of the output nodes becomes equal to the ratio of a desiredoutput amplitude value at the output nodes to the difference between thecharged potential VM of each of the capacitors 1a and 1b to theprecharged potential at the output nodes (i.e., about Vcc/2). Forexample, if the desired output amplitude value at the output nodes isVcc/10, CV: CD becomes equal to 2:10.

(Variation of Third Embodiment)

FIG. 9 shows a variation of the third embodiment, which is differentfrom the above semiconductor integrated circuit of FIG. 7 in that,instead of providing the two capacitors 1a and 1b for storing the higherpotential and the lower potential, respectively, attention is given tothe plate electrodes connected to the ground. As a result, the outputnode which shifts to the lower potential in the level-shifter on theupper stage and the output node which shifts to the higher potential inthe level-shifter on the lower stage are connected to the pair of plateelectrodes of the single capacitor 1, so that the respective potentialsare stored. Consequently, the number of capacitors can be reduced fromtwo to one, so that the effect of halving the layout area occupied bythe capacitor can be achieved.

(Fourth Embodiment)

FIG. 10 shows a fourth embodiment of the present invention. In the aboveembodiment shown in FIG. 7, the ratio of the capacitance value CV ofeach of the capacitors 1a and 1b to the parasitic capacitance value CDof the output node is set to 2:10. The present embodiment is constitutedso that the capacitance value CV of each of the capacitors 1a and 1b isreduced by setting the ratio at 1:10, thereby reducing the layout areaoccupied by each of the capacitors 1a and 1b. Specifically, it isarranged that the potential difference between each of the capacitors 1aand 1b and the output nodes prior to a short circuit is doubled fromabout Vcc/2 to about Vcc.

That is, in FIG. 10, a charge supplying means 700 enclosed in therectangle of broken lines has the two capacitors 1a and 1b and fourswitches SW1 to SW4. As shown in FIGS. 11 and 12, the above fourswitches SW1 to SW4 are controlled by a complementary pair of signalsWCLK and XWCLK which change in every cycle of the complementary pair ofclock signals CLK and XCLK. During a period T shown in FIG. 11, the twoswitches SW1 and SW3 are turned ON, while the other two switches SW2 andSW4 are turned OFF. Conversely, during the subsequent period δ, theswitches SW1 and SW3 are turned OFF, while the switches SW2 and SW4 areturned ON.

Consequently, during a period ε corresponding to a half of one cycle ofthe clock signal shown in FIG. 12, the output node Vout1 (which drops inpotential) is connected to the capacitor 1a by turning ON the PMOSFET 14controlled by, e.g., the input XVin1 of the level-shifter on the upperstage, so that the potential in the capacitor 1a is increased to thevicinity of the higher potential of the power-source line Vcc on thehigher potential side. On the other hand, the output node XVout2 (whichrises in potential) is connected to the capacitor 1b by turning ON theNMOSFET 15 controlled by, e.g., the input Vin2 of the level-shifter onthe lower stage, so that the potential in the capacitor 1b is lowered tothe vicinity of the lower potential of the power-source line Vss on thelower potential side. The state is maintained during a period ηcorresponding to the other half of one cycle of the clock signal.

Subsequently, the capacitor 1b is connected to the output node Vout1 orXVout1 which drops in potential in the level-shifter on the upper stageby switching the four switches SW1 to SW4, so that the potential in thecapacitor 1b is raised to the vicinity of the higher potential of thepower-source line Vcc on the higher potential side. On the other hand,the capacitor 1a is connected to the output node Vout2 or XVout2 whichrises in potential in the level-shifter on the lower stage, so that thepotential in the capacitor 1a is lowered to the vicinity of the lowerpotential of the power-source line Vss on the lower potential side.

Accordingly, the potential difference between each of the capacitors 1aand 1b and the output nodes connected thereto prior to a short circuitcan be set to about Vcc. As a result, the ratio of the capacitance valueCV of each of the capacitors 1a and 1b to the parasitic capacitancevalue CD of the output node can be reduced to 1:10, so that the layoutarea occupied by each of the capacitors 1a and 1b can be reduced.

(Variation of Fourth Embodiment)

FIG. 13 shows a variation of the above fourth embodiment, which uses avariation of the charge supplying means 700 of FIG. 10. A chargesupplying means 701 of FIG. 13 has a single capacitor 1 and two switchesSW1 and SW4 for connecting the capacitor 1 to the level-shifter on theupper stage and to the level-shifter on the lower stage. As shown inFIG. 14, during the period α corresponding to a half of one cycle of thecomplementary pair of clock signals CLK and XCLK, the switch SW1 isturned ON, while the switch SW4 is turned OFF. Conversely, during theperiod β corresponding to the other half of one cycle, the switch SW1 isturned OFF, while the switch SW4 is turned ON.

As shown in FIG. 15, the complementary pair of inputs Vin2 and XVin2 tothe level-shifter on the lower stage are modulated to have waveformswhich are phase shifted by half a cycle from those of the complementarypair of inputs Vin1 and XVin1 to the level-shifter on the upper stage.

Consequently, as shown in FIG. 15, when the switch SW1 is turned ONduring the period α corresponding to a half of one cycle of the clocksignal, the capacitor 1 is connected to that one of the complementarypair of output nodes Vout1 and XVout1 (Vout1 in the drawing) which dropsin potential in the level-shifter on the upper stage, so that thecapacitor 1 is charged to the vicinity of the higher potential of thepower-source line Vcc. Subsequently, when the switch SW4 is turned ONduring the period β corresponding to the other half of one cycle of theclock signal, the capacitor 1 is connected to that one of thecomplementary pair of output nodes Vout2 and XVout2 (XVout2 in thedrawing) which rises in potential in the level-shifter on the lowerstage. As a result, the charge in the capacitor 1 is released to theoutput node XVout2, so that the potential in the capacitor 1 is loweredto the vicinity of the lower potential of the power-source line Vss.

Thus, in the present variation, the single capacitor 1 is alternatelyconnected to the output node of a potential in the vicinity of thehigher potential Vcc and to the output node of a potential in thevicinity of the lower potential Vss in every half cycle of the clocksignal. Consequently, the ratio of the capacitance value CV of thecapacitor 1 to the parasitic capacitance value CD of the output node canbe reduced to 1:10, so that the layout area occupied by the capacitorcan be reduced.

(Fifth Embodiment)

Next, a semiconductor integrated circuit according to a fifth embodimentof the present invention will be described with reference to FIG. 16.

In the semiconductor integrated circuit of FIG. 16, three differentcomplementary-type level-shifters are connected in series between thepower source Vcc and the ground line Vss.

The level-shifter on the middle stage is shown in FIG. 3. Thelevel-shifter on the upper stage is shown in FIG. 6(a). Thelevel-shifter on the lower stage is shown in FIG. 6(b). However, all theMOSFETs of the level-shifter on the middle stage are composed ofNMOSFETs 24, 25, and 26. All the MOSFETs of the level-shifter on theupper stage are composed of PMOSFETs 14, 16, and 17. All the MOSFETs ofthe level-shifter on the lower stage are composed of NMOSFETs 13, 15,and 18.

The capacitor 1a of the level-shifter on the upper stage and thecapacitor 1b, which is one of the two capacitors 1b and 1c of thelevel-shifter on the middle stage, are connected to a power-source line22 of a higher potential VU via three NMOSFETs 21. The other capacitor1c of the level-shifter on the middle stage and the capacitor 1d of thelevel-shifter on the lower stage are connected to a power-source line 23of a lower potential VL via three NMOSFETs 27.

The higher potential VU on the above power-source line 22 is set to3(Vcc-Vss)/4, while the lower potential VL of the power-source line 23is set to (Vcc-Vss)/4.

Each of the PMOSFET 17 of the above level-shifter on the upper stage,the NMOSFET 24 of the above level-shifter on the middle stage, and theNMOSFET 18 of the above level-shifter on the lower stage functions as aprecharging means for short-circuiting the two corresponding outputnodes and precharging them to equal potentials. The four capacitors 1ato 1d constitute a charge accumulating means. Each of the power-sourcelines VU and VL of an intermediate potential functions as a chargingmeans for charging each of the above capacitors 1a to 1d to theintermediate potentials VU and VL (potentials different from the aboveprecharged potentials at the output nodes). These precharging means,charge accumulating means, and charging means constitute a chargevariation equalizing means.

On the other hand, the PMOSFET 14 of the level-shifter on the upperstage, the NMOSFET 25 of the level-shifter on the middle stage, and theNMOSFET 15 of the level-shifter on the lower stage constitute a chargeredistributing means for connecting the capacitors 1a to 1d to thecorresponding output nodes Vout1, XVout1, Vout2, XVout2, Vout3, andXVout3, respectively, and redistributing the charge accumulated in thecapacitors 1a to 1d to the output nodes.

Next, a description will be given to a control method of thesemiconductor integrated circuit of the present embodiment. As shown inFIG. 18, the level-shifter on the middle stage is operated insynchronization with the complementary pair of clock signals CLK andXCLK. During the period α corresponding to a half of one cycle of theclock signals, the NMOSFET 24 is turned ON so as to short-circuit thecomplementary pair of outputs VOUT2 and XVOUT2, thereby precharging eachof the outputs to an approximately intermediate potential (1/2 Vcc=Vc)between the power-source line 23 at the lower potential and thepower-source line 22 at the higher potential. Meanwhile, each of thecapacitors 1b and 1c is charged to the higher potential of thepower-source line 22 and to the lower potential of the power-source line23, respectively. During the subsequent period β corresponding to theother half of one cycle, the nodes of the above capacitors 1b and 1c areconnected to the complementary pair of outputs Vout2 and XVout2 via theNMOSFEs 25 and 26 which are turned ON by the complementary pair ofinputs Vin2 and XVin2, respectively.

Similarly in the level-shifter on the upper stage, during the period αcorresponding to a half of one cycle of the clock signals, thecomplementary pair of outputs Vout1 and XVout1 are precharged to theintermediate potentials of the respective potentials obtained when theyare complementarily outputted, while the capacitor 1a is charged to thepotential VU of the power-source line 22 of the higher potential.Subsequently, during the period β corresponding to the other half of onecycle, the node A of the above capacitor 1a and the power-source lineVcc of the higher potential are connected to the above complementarypair of outputs Vout1 and XVout1 via the PMOSFETs 16 and 14 which havebeen turned ON by the above complementary pair of inputs Vin1 and XVin1.

Similarly in the level-shifter on the lower stage, during the period αcorresponding to a half of one cycle of the clock signals, thecomplementary pair of outputs Vout3 and XVout3 are precharged to theintermediate potentials of the respective potentials obtained when theyare complementarily outputted, while the capacitor 1d is charged to thepotential VL of the power-source line 23 of the lower potential.Subsequently, during the period β corresponding to the other half of onecycle, the node D of the above capacitor 1d and the power-source lineVss of the lower potential are connected to the above complementary pairof outputs Vout3 and XVout3 via the NMOSFETs 13 and 15 which have beenturned ON by the above complementary pair of inputs Vin3 and XVin3.

As shown in FIG. 17, the above complementary pair of inputs Vin2 andXVin2 of the level-shifter on the middle stage and the abovecomplementary pair of inputs Vin3 and XVin3 of the level-shifter on thelower stage are set on standby on the level of the ground potential Vssso that, after the inputs are determined, either one of the inputsshifts to the higher potential Vcc. On the other hand, the complementarypair of inputs Vin1 and XVin1 of the level-shifter on the upper stageare set on standby on the level of the higher potential Vcc so that,after the inputs are determined, either one of the inputs shifts to theground potential Vss. With the settings, the power-source line Vcc ofthe higher potential, the power-source lines VU and VL of the twointermediate potentials, and the power-source line Vss of the lowerpotential constitute a through-current preventing means for preventingthe flow of a through current resulting from a short circuit.

Alternatively, the above through-current preventing means can also beconstituted as shown in FIG. 17, so that MOSFETs 21, e.g., areinterposed between the points A and A', the points B and B', the pointsC and C', and the points D and D', which are shown in FIG. 16. In thiscase, the complementary pair of clock signals CLK and XCLK (signalsdifferent from the complementary pairs of inputs Vin1 and XVin1, Vin2and XVin2, and Vin3 and XVin3) control the through-current preventingmeans so that it is not turned ON until either one of each complementarypair of inputs shifts to a potential lower than the threshold voltage ofthe corresponding input NMOSFET.

Thus, in the present embodiment, during the period β corresponding tothe other half of one cycle of the clock signal, the charge abandoned bythe output node Vout1 or XVout1 which drops in potential in thelevel-shifter on the upper stage is used to raise the potential at theoutput node Vout2 or XVout2 which rises in potential in thelevel-shifter on the middle stage, as shown in FIG. 18. On the otherhand, the charge released by the output node Vout2 or XVout2 which dropsin potential in the level-shifter on the middle stage is used to raisethe potential at the output node Vout3 or XVout3 which rises inpotential in the level-shifter on the lower stage. Consequently, onlythe level-shifter on the upper stage should be supplied with the chargefrom the outside, while the other two level-shifters do not require thesupply of additional charge, which enables a reduction in powerconsumption.

Below, a description will be given to the case where n level-shiftersare assumedly operated in each of the present and conventionalembodiments so as to compare the degree of reduction in powerconsumption attained by the present embodiment with that attained by theconventional embodiments.

In the case where the voltage level is lowered by means of the internalpower-source circuit as in the first conventional embodiment (JapaneseLaid-Open Patent Publication No. 4-211515), if the power-source voltage,the output current, and the voltage after the voltage drop arerepresented by Vcc, IH, and VH, respectively, power consumption in theinternal power-source circuit becomes n·IH·(Vcc-VH), while powerconsumption in the n circuits becomes n·IH·VH, so that the total powerconsumption amounts to Ptotal=n·IH·Vcc.

In the case where the voltage level is reduced by operating theadditional n circuits in series as in the second conventional embodiment(Japanese Laid-Open Patent Publication No. 4-302463), the voltageapplied to each of the circuits becomes VCC/n, so that the total powerconsumption Ptotal becomes Ptotal=(VCC/n·IH)·n+α=IH·Vcc+α. Here, thepower consumption α accounts for the through current allowed to flowwhen the voltage (Vcc-Vss) is divided with resistors. Since the throughcurrent IH always flows uselessly in series from the power source to theground terminal, it can be represented by IH·Vcc. Therefore, the totalpower consumption Ptotal becomes Ptotal=2·IH·Vcc.

In contrast, as shown in the equivalent circuit of FIG. 19, the presentinvention uses the n (in the drawing, n=3) level-shifters 501 in andfrom which equal charges are accumulated and released, i.e., in whichthe capacitors varying in potential have equal capacitances. Theselevel-shifters 501 are connected to each other only when the outputstherefrom have high impedance, so that if the capacitor of thelevel-shifter placed on the uppermost stage is solely charged, the otherlevel-shifters are operated by reusing the charge from their respectiveone-stage upper level-shifters, so that the total power consumptionPtotal becomes Ptotal=IH·Vcc.

Thus, as shown in FIG. 20, the power consumption in the presentinvention becomes half the power consumption in the case where the threelevel-shifters operate in series, as in the second conventionalembodiment, accompanied by the through current. In the presentinvention, there can also be achieved a reduction in power consumptionto 1/n, compared with the power consumption in the case where thevoltage is reduced by means of the internal power-source circuit as inthe first conventional embodiment.

Although the present embodiment has used the three differentcomplementary-type level-shifters in combination, if a few morelevel-shifters of the type disposed on the middle stage of FIG. 16 areconnected in series, it is possible to dispose four or more stages inseries. In this case, the level-shifter which outputs a potential higherthan the intermediate value ((Vcc+Vss)/2) between the higher potentialon the external power-source line Vcc and the lower potential on theexternal power-source line Vss from the output node consists ofPMOSFETs, while the level-shifter which outputs a potential lower thanthe intermediate value consists of NMOSFETs. With the structures, thegate/source voltages in the MOSFETs become sufficiently large, so thatstable operation can be performed.

Next, a circuit for generating the intermediate potentials VU and VLmentioned above is shown in FIG. 21. The intermediate potentialgenerator of FIG. 21 consists of current-mirror-type comparators 28, 29,and 30 which use the potential obtained through the resistive divisionby the four resistors R1, R2, R3, and R4 as the reference potential andoutput MOSFETs 31 to 33 controlled by the outputs from the comparators28 to 30.

The above intermediate potential generator is basically the same as thepower-source circuit of FIG. 28 shown in the conventional embodiment,except that it has a charge supplying ability greatly different fromthat of the conventional embodiment and that the supply of a new chargeis no more necessary since the charge is reused in the abovelevel-shifters on the intermediate and lower stages. Briefly, it is notnecessary to provide the intermediate potential generator shown in FIG.21 so that it performs a charge supplying function. The intermediatepotential generator is needed only when the first operating point isdetermined on turning ON the power source. As a result, it is no morenecessary for the output to have a low resistance as in the conventionalembodiment. It is possible to reduce in size the device constituting theintermediate potential generator of FIG. 21, so that the sum of thethrough currents IDC2, IDC3, IDC4, and IDC5 shown in the drawing can bereduced to the order of microamperes or less. It is also possible, asshown in FIG. 21, to further suppress the above through current byoperating the intermediate potential generator only on turning ON thepower source or only during the period corresponding to a half of onecycle of the clock signal CLK during which the clock signal CLK is Low,for example, and by halting its operation at all other times.

(Sixth Embodiment)

Next, a semiconductor integrated circuit according to a sixth embodimentof the present invention will be described with reference to FIG. 22.

In the present embodiment, an additional charge supplying means asdescribed above is not provided. Instead, the parasitic capacitance ofthe output node of a level-shifter is used as a charge supplying meansso that the charge accumulated at the output node is reused by anotherlevel-shifter.

That is, as shown in FIG. 22, the complementary-type level-shifter onthe upper stage has the same structure as that of the level-shiftershown in FIG. 6(a) and the complementary-type level-shifter on the lowerstage has the same structure as that of the level-shifter shown in FIG.6(b). All the MOSFETs of the level-shifter on the upper stage arecomposed of PMOSFETs 16, 14, and 17, while all the MOSFETs of thelevel-shifter on the lower stage are composed of NMOSFETs 13, 15, and18.

The midpoint (point A in the drawing) between the two PMOSFETs 14 of theabove level-shifter on the upper stage and the midpoint (point B in thedrawing) between the two NMOSFETs 15 of the above level-shifter on thelower stage are connected to the power-source line VM of theintermediate potential. The intermediate potential VM on the abovepower-source line VM is set at (Vcc-Vss)/2.

The PMOSFETs 14 of the above level-shifter on the upper stage and theNMOSFETs 15 of the above level-shifter on the lower stage constitute thecharge redistributing means for connecting and short-circuiting theoutput node Vout1 or XVout1 (Vout2 or XVout2) which shifts to the lowerpotential in one of the level-shifters to the output node Vout2 orXVout2 (Vout1 or XVout1) which shifts to the higher potential in theother level-shifter and thereby redistributing the charge released bythe output node which shifts to the lower potential to the output nodewhich shifts to the above higher potential. As for the other structure,it is the same as that of FIG. 7, so that the description thereof willbe omitted.

Next, a description will be given to a control method of thesemiconductor integrated circuit of the present embodiment. As shown inFIG. 24, during the period α corresponding to a half of one cycle of theclock signal CLK, the turning ON of the PMOSFET 17 and NMOSFET 18precharges the complementary pair of output nodes Vout1 and XVout1 andthe complementary pair of output nodes Vout2 and XVout2 of therespective level-shifters to the respective intermediate potentials,i.e., to the potentials of 3/4 Vcc and 1/4 Vcc.

Subsequently, during the period β corresponding to the other half of onecycle of the clock signal CLK, that one of the outputs nodes Vout1 andXVout1 which drops in potential in the level-shifter on the upper stagehaving high impedance as a result of turning OFF the PMOSFET 17 and thatone of the outputs nodes Vout2 and XVout2 which rises in potential inthe level-shifter on the lower stage are short-circuited via the PMOSFET14 which has been turned ON by either of the complementary pair ofinputs Vin1 and XVin1 and via the NMOSFET 15 which has been turned ON byeither of the complementary pair of inputs Vin2 and XVin2, respectively,and set at the intermediate potential of the power-source line Vc. Onthe other hand, the other output node Vout1 or XVout1 of the abovelevel-shifter on the upper stage is connected to the power-source lineVcc of the higher potential, while the other output node Vout2 or XVout2of the level-shifter on the lower stage is connected to the power-sourceline Vss of the lower potential.

As shown in FIG. 24, the above complementary pair of inputs Vin1 andXVin1 to the level-shifter on the upper stage are set on standby on thelevel of the ground potential Vss so that, after the inputs aredetermined, either one of the inputs solely shifts to the lowerpotential Vss. On the other hand, the complementary pair of inputs Vin2and XVin2 to the level-shifter on the lower stage are set on standby onthe level of the lower potential Vss so that, after the inputs aredetermined, either one of the inputs solely shifts to the higherpotential Vcc. With the settings, the flow of a through currentresulting from a short circuit between the power-source line Vcc of thehigher potential and the power-source line Vss of the lower potential isprevented.

Alternatively, the above semiconductor integrated circuit can also beconstituted as shown in FIG. 23, so that MOSFETs 50, 51, and 52, e.g.,are provided between the nodes A and B, between the node E and thepower-source line Vcc, and between the node F and the ground powersource Vss, which are shown in FIG. 22. With the constitution, thesemiconductor integrated circuit of the present embodiment is turned ONonly when either one of each complementary pair of inputs shifts to thethreshold voltage of the input MOSFET.

Thus, as can be seen from FIG. 24, the charge released by that one ofthe output nodes Vout1 and XVout1 which drops in potential in thelevel-shifter on the upper stage can be reused by that one of the outputnodes Vout2 and XVout2 which rises in potential in the level-shifter onthe lower stage, so that power consumption can be reduced in the presentembodiment.

(Seventh Embodiment)

Next, a seventh embodiment of the present invention will be describedwith reference to FIG. 25.

The present embodiment was obtained by developing the above sixthembodiment, in which two level-shifters of the type shown in FIG. 22 areprovided on the upper stage and two level-shifters of the type shown inFIG. 22 are provided on the lower stage. The four level-shifters intotal constitute the semiconductor integrated circuit.

Between the level-shifter on the first stage and the level-shifter onthe second stage is disposed a power-source line 55 of the firstintermediate potential VU (VU=3(Vc-Vss)/4). Between the level-shifter onthe second stage and the level-shifter on the third stage is disposed apower-source line 56 of the second intermediate potential Vc(Vc=2(Vc-Vss)/4). Between the level-shifter on the third stage and thelevel-shifter on the fourth stage is disposed the power-source line 57of the third intermediate potential VL (VL=(Vc-Vss)/4).

As for the other structure, it is the same as that of FIG. 22, so thatthe description thereof will be omitted.

In the present embodiment, as shown in FIG. 26, the four pairs of outputlines having the amplitude of Vcc/4 are operated by mutually reusing acharge. Consequently, of the four level-shifters, only the level-shifteron the uppermost stage connected closest to the power source is suppliedwith a charge required for a potential shift of Vcc/8 from the externalpower-source line Vcc, while the other three level-shifters operate withthe reused charge.

The above sixth and seventh embodiments can be made the best use of in acircuit in which the output nodes (output lines) have equal parasiticcapacitances and a potential for turning ON either one of thetransistors to which a complementary pair of inputs are connected can beprovided comparatively easily. For example, in a memory circuit operablewith the synchronous signal, if a data transfer system circuitoriginally operating with 16, 32, 64, or 128 bits is substituted by adata transfer circuit operating with 4 or 8 bits, power consumption inthe memory can be reduced.

(Eighth Embodiment)

FIG. 31 shows an eighth embodiment of the present invention. Thesemiconductor integrated circuit shown in the drawing comprises aplurality of level-shifters each including, e.g., the above dischargecircuit 9' and precharging means 11' of FIG. 4, the principal portionsof which are shown in blocks. As shown in FIG. 32, each block has a pairof output elements IDi and XIDi, a pair of output elements Di and XDi,and level setting elements H and L for each pair of outputs. In thedrawings, F represents a functional circuit, which is specifically aswitch composed of a transistor. Here, by way of example, let attentionbe given to the output D. If the output element Di is higher in levelthan the output element XDi in the same pair, the functional circuit Fperforms the function of connecting the output D to the level settingelement H. If the output element Di is lower in level than the outputelement XDi, on the other hand, the functional circuit F connects theoutput D to the level setting element L. In the drawings, EQ representsa signal for equalizing a pair of output lines. FIG. 33 shows theoperating tables of the respective blocks.

The above pair of input elements IDi and XIDi in each block correspondto the complementary pair of inputs Vin and XVin of FIG. 25. The pair ofoutput elements Di and XDi correspond to the complementary pair ofoutput nodes Vout and XVout of FIG. 25. The level setting elements H andL correspond to the intermediate potentials VU, VC, and VL of FIG. 25.The functional circuit F corresponds to the MOSFETs 14 and 15 of FIG.25.

In the above seventh embodiment of FIG. 25, the plurality oflevel-shifters are connected between the power source Vcc and the groundline Vss, as diagrammatically shown in FIG. 34. During the first periodof one cycle of the system clock, one of the pair of output lines of alevel-shifter and one of the pair of output lines of an adjacentlevel-shifter are short-circuited and, during the subsequent secondperiod, the other output lines of the respective level-shifters areshort-circuited, so that the above operation in one cycle is repeatedlyperformed. In the present embodiment, in contrast, a plurality of groupseach consisting of the respective blocks connected in series areprovided (two groups in FIGS. 31 and 35), as shown in FIG. 35, so thatthese groups and a single level-shifter are connected between the powersource Vcc and the ground line Vss.

In the structure of the present embodiment, since the plurality ofblocks are connected in parallel, the total capacitance of thecapacitances Cd0 to Cdn of the pairs of the output lines in therespective blocks becomes n×Cdn, as shown in the equivalent circuit ofFIG. 35. On the other hand, the total Capacitance of the group with noparallel connection becomes 1×Cdn. Accordingly, in the structure of FIG.31, the potential of the pair of outputs of each block is determined bythe capacitance ratio among the groups and, as shown in FIGS. 36(a) and36(b), an amplitude 70 of 1:n is obtained. Consequently, the blocksconnected in parallel operate with a small amplitude, while theindividual blocks which are not connected in parallel operate with alarge amplitude.

In the above third embodiment shown in FIGS. 7 and 8, it was necessaryto additionally provide a capacitance smaller than that of the pair ofoutput lines through layout or process. In the present embodiment,however, it is sufficient to increase the number of blocks connected inseries in order to produce a small capacitance. Conversely, in order toproduce a large capacitance, it is sufficient to increase the number ofblocks connected in parallel. Briefly, it is possible to control theamplitude distribution of the outputs by electrically controlling theeffective capacitance.

(Variation of Eighth Embodiment)

FIGS. 37 and 38 show a variation of the eighth embodiment. In the aboveeighth embodiment, the two groups are disposed in the upper and lowerpositions with the single block interposed therebetween. Conversely, thepresent embodiment is constituted so that two independent blocks aredisposed in the upper and lower positions with a group interposedtherebetween. The waveform of FIG. 6(b) represents the operation of thevariation.

(Ninth Embodiment)

FIG. 39 shows a ninth embodiment of the present invention. In thedrawing, a plurality of blocks are connected in series and first,second, and third down converters 101 to 103 are disposed in the chipbetween the external power source Vcc and the potential Vss of theground line. An internal power-source line 104 of the higher potentialVup is withdrawn from between the first and second down converters 101and 102. An internal power-source line 105 of the lower potential Vlw iswithdrawn from between the second and third down converters 102 and 103.Between the internal power-source lines 104 and 105 are disposed twodecoupling capacitors 106 each having a comparatively large capacitance.As shown in FIG. 40, however, the input to the circuit has a fullamplitude (Vcc-Vss), while the output amplitude has a potential obtainedby equally dividing the potential between the higher potential Vup andthe lower potential Vlw by a plural number (six in the drawing).

In the present embodiment, the higher potential Vup and lower potentialVlw internally generated by means of the three down converters 101 to103 are stabilized by the decoupling capacitors 106 each having a largecapacitance, so that, compared with the case where the plurality ofblocks in series are connected directly between the external powersource and the ground line, the problem of bounce of the potential inthe external power source and of the potential on the ground line due tonoise can be solved more satisfactorily.

Although the present embodiment has used the three down converters 101to 103 to reduce the external power-source voltage, it will easily beappreciated that the present invention can similarly be applied to thecase where the external power-source voltage is raised so that theresulting voltage is used as the internal power-source voltage.

(Variation of Ninth Embodiment)

FIG. 56 shows a fist variation of the ninth embodiment of the presentinvention. Unlike the above ninth embodiment in which the two decouplingcapacitors 106 are interposed between the internal power-source line 104of the higher potential Vup and the internal power-source line 105 ofthe lower potential Vlw, the present variation is constituted so that,in the case of connecting the external power-source line and the groundline directly to a plurality of blocks connected in series, decouplingcapacitors 106' which are equal in number to the blocks connected inseries are connected directly between the above external power-sourceline and the ground line, thereby connecting the decoupling capacitors106' to the nodes A, B, . . . C between the respective blocks.

Thus, as shown in FIG. 57, even when noises having values of ΔV1 and ΔV2occur in the voltage Vcc of the external power source in the presentembodiment, the voltages of the noises are divided by the decouplingcapacitors 106', so that only noises having lower values of ΔV1/n andΔV2/n obtained through voltage division by the number of the couplingcapacitors 106' . . . affect the respective blocks, so that the problemof bounce of the potential in the external power source and of thepotential on the ground line due to noises can be solved moresatisfactorily.

(Second Variation of Ninth Embodiment)

FIG. 58 shows a second variation of a ninth embodiment of the presentinvention. Unlike the above ninth embodiment in which the influence of anoise in the external power source is suppressed by the decouplingcapacitors 106, the present variation is constituted so that a pluralityof blocks in series are connected directly to the external power sourceand to the ground line, but the outputs from the blocks affected bynoises in the external power source and on the ground line are not usedas signals.

That is, in FIG. 58, the number of the blocks connected in series is setto 10, which is larger than the number (e.g., 8 bits) of required setsof parallel data by 2. Of the ten blocks, the output from the block onthe uppermost stage to which the power source Vcc is applied and theoutput from the block on the lowermost stage connected to the groundline of the voltage Vs(0 V) are not used as the above parallel data.

Thus, in the present variation, even when noises occur in thepower-source voltage Vcc and in the ground-line voltage Vss as shown inFIG. 59, only the block on the uppermost stage and the block on thelowermost stage are directly influenced, while the outputs from theintermediate blocks are not influenced by the above noises.Consequently, the parallel data can be outputted without being affectedby the noises.

(Tenth Embodiment)

FIG. 41 shows a tenth embodiment of the present invention. The tenthembodiment in the drawing is constituted so that, in the case where a 0bit (the least significant bit (LSB)) to the most significant bit (MSB)are allocated to the plurality of level-shifters connected in seriesbetween the power source Vcc and the ground line (first power source andsecond power source), the least significant bit LSB which varies mostfrequently is allocated to the level-shifter on the uppermost stage fromwhich the output line is connected directly to the power source line.

Thus, as shown in FIGS. 42 and 43, the present embodiment is constitutedso that, as the level-shifter has its output potential in a positionfurther away from the power source Vcc, the substrate bias voltage Vbs(Vbs=Vcc-Vb) of the two NMOSFETs 14 constituting the chargeredistributing means in FIG. 25 becomes larger. Accordingly, thesubstrate bias voltage becomes larger with respect to the waveform 3than with respect to the waveform 1, as shown in FIG. 42. On the otherhand, the level-shifter constituting the least significant bit LSB isconnected directly to the power source line in consideration of theincreasing operation delay time of the MOSFET 14 as shown in FIG. 43(b),thus minimizing the delay time of the level-shifter on the uppermoststage of all the level-shifters.

Thus, in the present embodiment, in the case of incrementing an address,e.g., the least significant bit varies most frequently, while the upperbits vary less frequently. However, since stable operation of thelevel-shifter in the least significant bit is ensured, the entirecircuit can operate more stably.

(Eleventh Embodiment)

FIG. 44 shows an eleventh embodiment of the present invention. Thestructure shown in the drawing is used in the case where the memory isconnected to a graphic controller with N bit lines. For example, verymany bits of 512 bits are divided into 64 groups of 8 bits. In thiscase, when one set of data is represented in 8 digits (8 bits), the samedigits of four sets of data form one group.

FIG. 45 shows a frequency divider used to drive the above very manybits. The frequency divider of FIG. 45 consists of a plurality of (fourin the drawing) D-latch circuits 107 in cascade and produces outputs ofOT-1, OT-2, OT-3, and OT-4 which are obtained by multiplying thefrequency of the input IN by power-of-2 numbers on the respectivestages. The output OT-1 from the above frequency divider is used as anoperating signal for the group consisting of eight least significantbits LSB. The output OT-2 is used as an operating signal for the groupconsisting of eight bits in positions one-bit higher than the leastsignificant bits LSB. Likewise, the output OT-3 is used as an operatingsignal for the group consisting of eight bits in positions two-bithigher than the least significant bits LSB and the output OT-4 is usedas an operating signal for the group consisting of eight bits inpositions three-bit higher than the least significant bits LSB.

Thus, in the present embodiment, each group has a different operatingfrequency. Since the group consisting of the most significant bitsoperates only with a frequency which is 1/64 the system clock, the ratioof the power consumption of the present embodiment to that of the priorart is increased to n:2 as the number n of groups increases.Consequently, power consumption can be reduced significantly.

Although each of the groups includes the same number of bits in thepresent embodiment, different groups may include different numbers ofbits.

(Twelfth Embodiment)

FIG. 48 shows a twelfth embodiment of the present invention. In thedrawing, the above clock CLK of FIG. 25 is used as a first clock CLK1and a P-type MOSFET 121 is additionally disposed between the P-typeMOSFET 16 and the power source Vcc, so that the above MOSFET 121 iscontrolled by a second clock CLK2 via an inverter 122. As for the otherstructure, it is the same as that of FIG. 25, so that the descriptionthereof will be omitted by providing like reference numerals to likecomponents.

The above second clock CLK2 has a frequency double the frequency of theabove first clock CLK1. As a structure for generating the above secondclock CLK2, a frequency divider having the D-latch circuit 107 shown inFIG. 45 is used.

Thus, in the present embodiment, a charge Q1 from the power source Vccis not supplied every cycle to the level-shifter which is connectedclosest to the power source Vcc as shown in FIG. 50, but a charge Q2 issupplied from the power source every two cycle. The charge Q2 suppliedfrom the power source Vcc is represented by hatched portions in FIGS. 49and 50. As a result, the level-shifter is connected to the power sourceVcc during the period of t=T1, while it is disconnected from the powersource Vcc during the subsequent period of t=T2 as shown in FIGS. 49 and50, so that the total amount of charge during the two periods isrepresented by the following inequality:

    (Q1+Q1)>Q2+"0"

where Q2=1.5×Q1.

As can be seen from the above inequality, a reduction in powerconsumption can be achieved in the present embodiment. It will easily beappreciated from the operating waveform that the amplitude value isreduced when a charge is not supplied. However, since a pair ofdifferential output lines are used, a high margin can be ensured.

(Thirteenth Embodiment)

FIGS. 51 and 52 show a thirteenth embodiment of the present invention.Unlike the above semiconductor integrated circuit of FIG. 25 in whichthe four level-shifters are disposed in series between the power sourceand the ground line, eight level-shifters are disposed in series in thesemiconductor integrated circuit of the present embodiment, as shown inFIGS. 51 and 52. As described in the above seventh embodiment of FIG.25, the outputs from the respective level-shifters are different fromeach other and have eight reference levels in the range from thepotential level of the power source to the ground level. In FIG. 52, areference numeral 150 designates a potential detecting circuit(potential detecting means) composed of an N-type MOSFET to which thepair of output lines of the level-shifter serving as a driver areinputted and a reference numeral 151 designates a potential detectingcircuit (potential detecting means) composed of a P-type MOSFET. Theabove N-type potential detecting circuit 150 has an input gate element150a composed of an N-type MOSFET, as shown in FIG. 53. The P-typepotential detecting circuit 151 has an input gate element 151a composedof an N-type MOSFET, as shown in FIG. 54.

The above potential detecting circuit 150 composed of the N-type MOSFETdetects the input level of the precharged potential at the output nodeof the corresponding level-shifter which ranges from the intermediatepotential between the power-source potential Vcc and the groundpotential to the power-source potential Vcc. The input level rangingfrom the above intermediate potential to the ground potential isdetected by the potential detecting circuit 151 composed of the P-typeMOSFET. The output from the above potential detecting means 150 composedof the N-type MOSFET is connected to the respective source elements 150dof a pair of N-type transistors 150c of a CMOS-type flip-flop 150b. Onthe other hand, the output from the potential detecting circuit 151composed of the P-type MOSFET is connected to the respective sourceelements 151d of a pair of P-type transistors 151c of a CMOS-typeflip-flop 151b.

FIG. 55 shows a sensing delay in each of the above potential detectingcircuits composed of the N-type MOSFET and the P-type MOSFET withrespect to the input voltage. In the potential detecting circuit 150composed of the N-type MOSFET, the sensing delay does not present asignificant change with the input voltage ranging from the power-sourcevoltage Vcc to 1/2 Vcc. With the input voltage lower than 1/2 Vcc,however, the sensing delay increases abruptly, since the differencebetween the input voltage and the threshold voltage of the N-type MOSFETof the input element is reduced and hence the potential detectingcircuit 150 composed of the N-type MOSFET is turned ON onlyincompletely. Conversely, the potential detecting circuit 151 composedof the P-type MOSFET is completely turned ON with the input voltage inthe range of 1/2 Vcc to Vss.

Thus, the present embodiment can provide a potential detecting circuitwhich exhibits an excellent sensitivity around the output potential(reference level) of the corresponding level-shifter (driver) if used inthe range with a reduced sensing delay.

As can be seen from FIGS. 51 and 52, the level-shifter composed of theN-type MOSFET is used with the input potential in the range of Vss to1/2 Vcc and the level-shifter composed of the P-type MOSFEt is used withthe input potential in the range of 1/2 Vcc to Vcc, while the potentialdetecting circuit composed of the N-type MOSFET is used with the inputpotential in the range of 1/2 Vcc to Vcc and the potential detectingcircuit composed of the P-type MOSFET is used with the input potentialin the range of Vss to 1/2 Vcc, so that a complementary relationship isestablished therebetween.

We claim:
 1. A level-shifter which receives an input signal switchingbetween two different values and shifts the amplitude value of the inputsignal to another amplitude value so that the resulting signal isoutputted from a single output node, the level-shifter comprising:afirst charge supplying means to be precharged to a first potential; asecond charge supplying means to be precharged to a second potentialwhich is different from the first potential; and a discharging means forselecting either one of the first and second charge supplying means inaccordance with the input signal and releasing the charge accumulated inthe selected charge supplying means to the output node.
 2. Acomplementary-type level-shifter which receives an input signalswitching between two different values and shifts the amplitude value ofthe input signal to another amplitude value so that the resulting signalis outputted from a pair of output nodes, the level-shifter comprising:afirst charge supplying means to be precharged to a first potential; asecond charge supplying means to be precharged to a second potentialwhich is different from the first potential; and a discharging means forreleasing the charge accumulated in the first charge supplying means toeither one of the pair of output nodes, while releasing the chargeaccumulated in the second charge supplying means to the other outputnode, in accordance with the input signal.
 3. A complementary-typelevel-shifter according to claim 2, further comprising:a prechargecircuit for precharging the pair of output nodes to a third potentialintermediate between the first potential of the first charge supplyingmeans and the second potential of the second charge supplying means. 4.A control method of a level-shifter according to claim 3, comprising thestep of:precharging, during a first period of one cycle, the pair ofoutput nodes to the third potential by means of the precharge circuit;precharging, during a second period of the one cycle, the first chargesupplying means and the second charge supplying means to the firstpotential and to the second potential, respectively; and releasing,during a third period after the first and second periods of the onecycle, the charge accumulated in the first and second charge supplyingmeans to the pair of output node by means of the discharging means.
 5. Asemiconductor integrated circuit, comprising:a plurality ofcomplementary-type driver circuits each of which receives an inputsignal switching between two different values and each of which has apair of complementary output nodes; two first charge variationequalizing means for respectively equalizing the pair of complementaryoutput nodes in each of two complementary-type driver circuits out ofthe plural driver circuits; and a second charge variation equalizingmeans for equalizing an output node which is one of the pair of outputnodes of one of the two complementary-type driver circuits and isincreased in potential to an output node which is one of the pair ofoutput nodes of the other one of the two complementary-type drivercircuits and is decreased in potential.
 6. A semiconductor integratedcircuit according to claim 5, wherein the plural complementary-typedriver circuits are disposed in series between a first power source anda second power source having a potential different from the potential ofthe first power source.
 7. A control method of a semiconductorintegrated circuit having at least two complementary-type drivercircuits each of which includes a pair of complementary output nodes,and at least two charge accumulating means, comprising the stepsof:during a first period of one cycle, precharging the pair of outputnodes in each of the two driver circuits to respective set potentials;during a second period of the one cycle, charging the two chargeaccumulating means to a set potential between the precharged potentialsof the respective two driver circuits; and during a third period afterthe first and second periods of the one cycle, connecting one of thecharge accumulating means to an output node which shifts to a higherpotential in one of the two driver circuits, while connecting the othercharge accumulating means to an output node which shifts to a lowerpotential in the other driver circuits.
 8. A control method of asemiconductor integrated circuit according to claim 7, wherein the firstperiod and the second period are coincident with each other.
 9. Acontrol method of a semiconductor integrated circuit having a pluralityof complementary-type driver circuits each of which includes a pair ofcomplementary output nodes, comprising the steps of:during a specifiedperiod, precharging the pair of output nodes in each of two drivercircuits out of the plural driver circuits to set potentials,respectively; and during a period after the specified period, connectingan output node which shifts to a higher potential in one of the twodriver circuits to an output node which shifts to a lower potential inthe other driver circuit of the two driver circuits.
 10. A controlmethod of a semiconductor integrated circuit according to claim 9,whereinoutputs from the respective pair of output nodes of the pluralcomplementary-type driver circuits are transferred to the outside as twoor more sets of parallel data, and the two or more sets of parallel dataare transferred to the outside using their respective voltage rangesobtained by preliminarily dividing a finite potential difference betweena first power source and a second power source by a plural number.
 11. Asemiconductor integrated circuit comprising at least twocomplementary-type circuits each of which receives a signal andcomprises first and second output nodes whose outputs are complementaryto each other,wherein each of the two circuits has a high potential sideconnecting terminal and a low potential side connecting terminal, thehigh potential side connecting terminal of one of the circuits beingconnected to the low potential side connecting terminal of the othercircuit, each of the circuits is provided with a functional circuit forswitching, in accordance with the signal input to the correspondingcircuit, between a first state in which the first output node and thesecond output node are connected to the high potential side connectingterminal and the low potential side connecting terminal, respectively,and a second state in which the first output node and the second outputnode are connected to the low potential side connecting terminal and thehigh potential side connecting terminal, respectively.
 12. Asemiconductor integrated circuit according to claim 11, wherein each ofthe two circuits comprises a complementary-type level-shifter whichshifts an amplitude value of the received signal to obtain acomplementary signal of another amplitude value so that thecomplementary signal is output to the first and second output nodes ofeach of the two circuits.
 13. A semiconductor integrated circuitaccording to claims 11 or 12, wherein the function circuit correspondingto one of the two circuits connects, when one of the first and secondoutput nodes of one of the two circuits shifts to a higher potential inaccordance with the input signal of the corresponding circuit, the oneoutput node shifted to the higher potential to the high potential sideconnecting terminal, andthe function circuit corresponding to the othercircuit connects, when one of the first and second output nodes of theother of the two circuits shifts to a lower potential in accordance withthe input signal of the corresponding circuit, the one output nodeshifted to the lower potential to the low potential side connectingterminal.
 14. A semiconductor integrated circuit according to claims 11or 12, wherein the low potential side connecting terminal of one of thetwo circuits is connected to the high potential side connecting terminalof a different complementary-type circuit other than the two circuits,andthe high potential side connecting terminal of the other circuit ofthe two circuits is connected to the low potential side connectingterminal of a further different complementary-type circuit other thanthe two circuit and the different circuit.
 15. A semiconductorintegrated circuit according to claims 11 or 12, wherein the lowpotential side connecting terminal of one of the two circuits isconnected to a low voltage source, and the high potential sideconnecting terminal of the other one of the two circuits is connected toa high voltage source.
 16. A semiconductor integrated circuit accordingto claims 11 or 12, wherein each circuit is provided with an equalizingcircuit which receives an equalization signal and equalizes the firstand second output nodes of the corresponding circuit when theequalization signal is received.
 17. A control method of a semiconductorintegrated circuit having at least two complementary-type circuits whichreceives a signal and comprises first and second output nodes whoseoutputs are complementary to each other, comprising the stepsof:inputting a signal to each of the two complementary-type circuits,and switching, in accordance with the signals input to the two circuits,between a first state in which the first and second output nodes of oneof the two complementary-type circuits are connected to the first andsecond output nodes, respectively, of the other complementary-typecircuit, and a second state in which the first and second output nodesof one of the two complementary-type circuits are connected to thesecond and first output nodes, respectively, of the othercomplementary-type circuit.
 18. A control method of a semiconductorintegrated circuit according to claim 17, wherein each of the twocircuits comprises a complementary-type level-shifter which shift anamplitude value of the received signal to obtain a complementary signalof another amplitude value so that the complementary signal is outputfrom the first and second output nodes of each of the two circuits. 19.A control method of a semiconductor integrated circuit according toclaims 17 or 18, wherein in the connection switching step, when apotential of one of the first and second output nodes of one of thecircuits is controlled to a higher potential and a potential of one ofthe first and second output nodes of the other circuit is controlled toa lower potential in accordance with the signals input to the twocircuits, the output node which shifts to the higher potential in theone circuit is connected to the output node which shifts to the lowerpotential in the other circuit.
 20. A control method of a semiconductorintegrated circuit according to claim 19, wherein in the connectionswitching step, when a potential of the other one of the first andsecond output nodes of the one of the circuits is controlled to thelower potential and a potential of the other one of the first and secondoutput nodes of the other circuit is controlled to the higher potentialin accordance with the signals input to the two circuits, the outputnode which shifts to the lower potential in the one circuit is connectedto the output node which shifts to the higher potential in a firstcomplementary-type circuit other than the two circuits while the outputnode which shifts to the higher potential in the other circuit isconnected to the output node which shifts to the lower potential in asecond complementary-type circuit.
 21. A control method of asemiconductor integrated circuit according to claim 19, wherein in theconnection switching step, when a potential of the other one of thefirst and second output nodes of one of the circuits is controlled tothe lower potential and a potential of the other one of the first andsecond output nodes of the other circuit is controlled to the higherpotential in accordance with the signals input to the two circuits, theoutput node of the one circuit which shifts to the lower potential isconnected to a low voltage source while the output node of the othercircuit which shifts to the higher potential is connected to a highvoltage source.
 22. A control method of a semiconductor integratedcircuit according to claims 17 or 18, further comprising the step ofequalizing the first and second output nodes of each of the circuitsupon equalization request other than that for controlling the respectivepotentials of the first and second output nodes of each circuit inaccordance with the input signal.